Filter circuit

ABSTRACT

A filter circuit whose cut-off frequency is easily controlled and which can be manufactured as an integrated circuit. The base electrode of a first transistor is connected to an input terminal by a reactance element, such as a capacitor. Second and third transistors are connected in differential amplifier configuration, and a current source is connected to the common connection of the emitter electrodes thereof. The emitter electrode of the first transistor is connected to the base electrode of the second transistor and the base electrode of the first transistor is connected to the collector electrode of the second transistor. An output terminal is connected to at least one of the collector and emitter electrodes of the first transistor.

BACKGROUND OF THE INVENTION

This invention relates to a filter circuit and, more particularly, to afilter circuit whose cut-off frequency is easily controlled and,moreover, which can be constructed as an integrated circuit.

It is conventional to fabricate filter circuits an integrated circuits.For cost effectiveness in manufacturing such integrated circuit filters,it is important that the number of external connections which are neededfor proper operation thereof, that is, the number of connectingterminals which must be provided, be minimized. Heretofore, activefilters have been manufactured as integrated circuits. However, intypical IC active filters, the resistance values of the variousresistive elements often cannot be as high as desired. This results in anon-uniform cut-off frequency. That is, in a particular run of IC activefilters, the cut-off frequency of one may differ from that of the other.

Another difficulty in IC active filters is that, since the temperaturecharacteristic of the resistive elements often is less thansatisfactory, the cut-off frequency of the filter is, to a significantextent, dependent upon temperature. Thus, the operating characteristicsof the IC active filter may become unstable with temperature deviations.

Yet another disadvantage in typical IC active filters is that a lowcut-off frequency for a high-pass filter is not easily obtainablebecause the resistance and capacitance values of the resistive andcapacitive elements therein are not as high as desired. That is,limitations on the resistive and capacitive values prevent the high-passfilter from having a relatively low cut-off frequency.

OBJECTS OF THE INVENTION

Therefore, it is an object of the present invention to provide animproved filter circuit which overcomes the aforenoted difficulties andproblems attending prior art IC active filters.

Another object of this invention is to provide a filter circuit whosecut-off frequency can be controlled easily, and which can bemanufactured as an integrated circuit.

A further object of this invention is to provide a variable filtercircuit whose cut-off frequency is controlled as a function of a controlsignal.

An additional object of this invention is to provide a filter circuitwhich exhibits a controllable cut-off frequency such that manufacturedfilters of different batches all can be controlled to exhibitsubstantially identical operating characteristics.

Yet another object of this invention is to provide an improved filtercircuit whose operation is relatively unaffected by temperature.

A still further object of this invention is to provide a high-passfilter circuit whose cut-off frequency may be made desirably low.

Various other objects, advantages and features of the present inventionwill become readily apparent from the ensuing detailed description, andthe novel features will be particularly pointed out in the appendedclaims.

SUMMARY OF THE INVENTION

In accordance with this invention, the filter circuit is provided with afirst transistor whose base electrode is connected to an input terminalby a reactance element, such as a capacitor. Second and thirdtransistors are connected in differential amplifier configuration andhave their emitter electrodes coupled in common to a current source. Theemitter and base electrodes of the first transistor are connected to thebase and collector electrode respectively, of the second transistor. Anoutput terminal is connected to at least one of the collector andemitter electrodes of the first transistor. Various embodiments of thefilter circuit are disclosed, including the use of the filter circuit ina controllable oscillator, as a phase shifter, and as a hue controlcircuit for a color television receiver.

BRIEF DESCRIPTION OF THE DRAWINGS

The following detailed description, given by way of example, will bestbe understood in conjunction with the accompanying drawings in which:

FIGS. 1-10 are schematic representations of various embodiments of thefilter circuit in accordance with the present invention.

DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS

Referring now to the drawings, wherein like reference numerals are usedthroughout, and in particular to FIG. 1, there is illustrated a basicembodiment of a filter circuit in accordance with the teachings of thepresent invention. The circuit shown herein is comprised of transistorsQ₁, Q₂ and Q₃. The base electrode of transistor Q₁ is connected to aninput terminal T₁ by a reactance element, shown herein as a capacitor C.It will be appreciated that, if desired, other reactive elements, suchas an inductance, may be used in place of capacitor C. The emitterelectrode of transistor Q₁ is connected to a reference potential, suchas ground, by a current source A₁. The emitter electrode also isconnected to an output terminal T₂.

Transistors Q₂ and Q₃ are connected in differential amplifierconfiguration with their emitter electrodes connected in common. Thecommon connection of these emitter electrodes is coupled to ground by acontrollable current source A₂. The base electrode of transistor Q₂ isconnected to the emitter electrode of transistor Q₁, and the collectorelectrode of transistor Q₂ is connected to the base electrode oftransistor Q₁. A current source A₃ also is connected to the collectorelectrode of transistor Q₂. A bias voltage source V_(BB) is connected tothe base electrode of transistor Q₃. Finally, the collector electrodesof transistors Q₁ and Q₃ are connected to a terminal T₃ supplied with anoperating potential +V_(cc).

In operation, let it be assumed that the input voltage applied toterminal T₁ is represented as V_(in), and let it be further assumed thatthe output voltage provided at the terminal T₂, that is, at the emitterelectrode of transistor Q₁, is represented as V_(out). With currentsource A₁ connected to the emitter electrode of transistor Q₁, it isappreciated that this transistor exhibits an emitter-followerconfiguration. Hence, if the base-emitter voltage drop across thistransistor is neglected, the voltage at the base electrode thereof issubstantially equal to the voltage at the emitter electrode thereof.Thus, the voltage provided at the base electrode of transistor Q₁ issubstantially equal to V_(out).

Let it be assumed that the angular frequency of the voltage provided atthe terminal T₁ is represented as ω. As a result of this signal, asignal current i_(s) flows through capacitor C. This signal current maybe represented as: ##EQU1##

It is seen that the voltage provided at the emitter electrode oftransistor Q₁ is applied to the base electrode of transistor Q₂. Thesignal current i_(s) flows through the collector-emitter circuit oftransistor Q₂ in response to this voltage V_(out). This signal currentalso flows through the collector-emitter circuit of transistor Q₃. Thissignal current may be determined by the voltage drop from the baseelectrode of transistor Q₂ to the base electrode of transistor Q₃. Sincethe voltage provided at the base electrode of transistor Q₃ is assumedherein to be the bias voltage V_(BB), it is appreciated that, for an ACsignal, the AC voltage across the base electrodes of transistors Q₂ andQ₃ is equal to V_(out). The resistance in this AC path traversed bysignal current i_(s) is equal to the emitter resistance of each oftransistors Q₂ and Q₃. If this emitter resistance is assumed to be equaland is represented as r_(e), then the signal current i.sub. s flowing inthe loop formed by transistor Q₂ and Q₃ is equal to the voltagedifference between the base electrode of transistor Q₂ and the baseelectrode of transistor Q₃, divided by the emitter resistance in thiscircuit. Thus, signal current i_(s) may be expressed as: ##EQU2##

The transfer function H (ω) of the filter shown in FIG. 1 is equal tothe output voltage divided by the input voltage. From equations (1) and(2), this transfer function may be expressed as: ##EQU3## The transferfunction expressed in equation (3) indicates that the filter circuit ofFIG. 1 functions as a high-pass filter whose cut-off frequency ω_(c) maybe represented as:

    ω.sub.c =1/2Cr.sub.e                                 (4)

Now, if the current which flows through current source A₂ is representedas 2I, then, since the emitter currents of transistors Q₂ and Q₃ areequal, the emitter current of each transistor is equal to I. The emitterresistance r_(e) is related to the emitter current I in each oftransistors Q₂ and Q₃ in accordance with the following expression:

    r.sub.e =kT/cI                                             (5),

wherein k is the Boltzmann constant, T is absolute temperature and q isthe charge of an electron.

If equation (5) is substituted into equations (3) and (4), then thetransfer function H(ω) and the filter cut-off frequency ω_(c) may berewritten as: ##EQU4##

It is, therefore, appreciated that the cut-off frequency of thehigh-pass filter shown in FIG. 1 can be changed, or controlled as afunction of the magnitude of the current flowing through current sourceA₂. Various embodiments of current source circuits are known to those ofordinary skill in the art, and an example thereof is described ingreater detail below. Since the current through a current source may becontrolled by a suitable control signal, it is further recognized thatthe operating characteristics, that is, the transfer function andcut-off frequency of the filter circuit illustrated in FIG. 1 may becontrolled in response to such a control signal. Consequently, since thecut-off frequency is established as a function of the current I, as isapparent from equation (7), and since this current can be controlledeasily, the cut-off frequency ω_(c) from one filter circuit to anothermay be suitably controlled so as to be uniform. Furthermore, even thoughequation (7) represents that the cut-off frequency is dependent upontemperature T, it is appreciated that this temperature dependency can becancelled by suitably controlling current I. For example, if thetemperature increases, the current may be increased, in response to theaforementioned control signal, so as to balance, or cancel, any effectupon the cut-off frequency attributed to this change in temperature.

Another advantage of the circuit shown in FIG. 1 is that, even thoughthe capacitance of capacitor C may be limited by reason ofstate-of-the-art integrated circuit fabrication techniques, the cut-offfrequency ω_(c) may be made as low as desired merely be reducing thecurrent I. Thus, it is seen that the embodiment shown in FIG. 1overcomes the aforenoted disadvantages of prior art IC active filtersand that the present invention is capable of providing high-pass filtersof uniform cut-off frequency, of good temperature immunity and of lowcut-off frequencies, as desired. Also, since the cut-off frequency isdetermined by the current of current source A₂, the cut-off frequencycan be varied rapidly over a relatively wide range merely by controllingthe current source. That is, as the current I changes, the cut-offfrequency changes in the corresponding manner.

Yet another advantage of the embodiment shown in FIG. 1 is that theoutput impedance thereof is determined by the output impedance oftransistor Q₁. It is appreciated that this output impedance isrelatively low so that the illustrated filter circuit can be connectedto other circuits without undesired leading effects. Hence, this circuitmay be incorporated easily in a multi-stage device.

The signal current i_(s) is, of course, a function of the input voltageV_(in). As this signal current varies, the emitter resistance r_(e) ofeach of transistors Q₂ and Q₃ also may vary. However, in view of thedifferential amplifier configuration of these transistors, it should berecognized that such variations in the emitter resistances thereof arein opposite directions. That is, if the emitter resistance of onetransistor increases by reason of signal current i_(s), the emitterresistance of the other transistor decreases. Consequently, such achange in the emitter resistances because of a change in the signalcurrent has a cancelling effect. As a result thereof, the apparentchange in emitter resistance r_(e) caused by signal current i_(s) isnegligible. This means that the illustrative filter circuit exhibits agood distortion factor and a wide dynamic range.

Another embodiment of the present invention is illustrated in FIG. 2.This embodiment differs from that shown in FIG. 1 in that current sourceA₃, which is coupled to the collector electrode of transistor Q₂, isconstituted by a transistor Q₆ which is connected to a transistor Q₅ ina current mirror circuit configuration. Also, in the embodiment of FIG.2, a particular example of current source A₂ is illustrated ascomprising transistor Q₄ whose base electrode is supplied with a controlvoltage E via a voltage divider circuit formed of resistors R₁ and R₂.Control voltage E is applied to a terminal T₄ which, in turn, isconnected to the voltage divider circuit.

When a current mirror circuit is used as current source A₃, as shown inFIG. 2, signal current i_(s) through capacitor C is doubled. That is,because of this current mirror circuit, signal current i_(s) exhibitstwice the value in FIG. 2 than in FIG. 1 for the same input voltageV_(in). Consequently, the cut-off frequency ω_(c) for the embodiment ofFIG. 2 is twice the cut-off frequency for the embodiment of FIG. 1, andmay be expressed as:

    ω.sub.C =q/kT·I/C                           (8).

Of course, if control voltage E is varied, the collector current oftransistor Q₄ is varied in response thereto. Hence, a change in thecontrol voltage results in a change in current I so as to vary or adjustthe cut-off frequency ω_(c).

From equation (8), it is seen that the cut-off frequency is dependentupon temperature T. For example, if the temperature increases, it wouldappear that the cut-off frequency ω_(c) decreases. However, in theembodiment of FIG. 2, since current source A₂ is constituted bytransistor Q₄, it is recognized that the increase in temperature Tresults in an increase in the collector current of transistor Q₄. Thismeans that, as temperature T changes, current I in equation (8) changesin a similar manner. Consequently, any dependency of the cut-offfrequency on temperature is cancelled. That is, a reduction in thecut-off frequency caused by an increase in temperature is cancelledbecause current I also increases by a corresponding amount. Thus, inpractice, the cut-off frequency ω_(c) is not temperature dependent toany significant extent and, therefore, the temperature characteristic ofthe illustrated filter circuit is improved over the prior art.

Another embodiment of the present invention is illustrated in FIG. 3.This embodiment differs from that described above with respect to FIG. 2in that current source A₂ is not specified as being of any specificconstruction, although it may, of course, be constructed as shown inFIG. 2; and in that a series of n diodes is connected in the emittercircuit of transistor Q₂ and a series of n diodes is connected in theemitter circuit of transistor Q₃. As shown specifically, diodes D₁₁,D₁₂, . . . D_(1n) are connected in series between the emitter electrodeof transistor Q₂ and current source A₂. Similarly, diodes D₂₁, D₂₂, . .. D_(2n) are connected in series between the emitter electrode oftransistor Q₃ and the current source. Each diode exhibits a resistancer_(e). That is, the resistance of each diode is equal to the emitterresistance of transistor Q₂ (and also transistor Q₃). This equalresistance may be attained easily in accordance with conventionalintegrated circuit manufacturing techniques. Hence, the effectiveresistance between the base electrode of transistor Q₂ and currentsource A₂ is equal to (n+1) r_(e). Similarly, the effective resistancebetween current source A₂ and the base electrode of transistor Q₃ isequal to (n+1) r_(e). Thus, in the embodiment shown in FIG. 3, theresistance in the path traversed by signal current i_(s) is equal to2(n+1) r_(e). With this value of resistance, the derivation of themathematical expression for cut-off frequency ω_(c) results in thefollowing: ##EQU5##

A comparison of equations (8) and (9) indicates that the cut-offfrequency ω_(c) for the embodiment of the filter circuit shown in FIG. 3will, for equal currents I and for equal capacitance C, be lower thanthe cut-off frequency for the embodiment of the filter circuit shown inFIG. 2. That is, the cut-off frequency for the embodiment shown in FIG.3 is 1/(n+1) the cut-off frequency for the embodiment of FIG. 2. It isappreciated that if the cut-off frequency is to be the same for bothembodiments, then the current I in equation (9) must be (n+1) times asgreat as the current I in equation (8). If the magnitude of the currentI relative to the magnitude of signal current i_(s) is large, as in theFIG. 3 embodiment, the dynamic range of the filter circuit is increased.Thus, for equal cut-off frequencies, the dynamic range of the embodimentshown in FIG. 3 is wider than the dynamic range of the embodiment shownin FIG. 2.

Referring now to the embodiment shown in FIG. 4, it is recognized thatthis embodiment is substantially similar to that described above withrespect to FIG. 3, except that the diode resistances (n+1) r_(e) of theFIG. 3 embodiment are replaced by ohmic resistances R₃ and R₄ in theFIG. 4 embodiment. Hence, the FIG. 4 embodiment attains substantiallythe same results and exhibits substantially the same effects as the FIG.3 embodiment. That is, for the same value of current I, and for the samevalue of capacitance C, the cut-off frequency for the embodiment of FIG.4 is substantially reduced relative to the cut-off frequency for theembodiment of FIG. 2, dependent upon the values of resistances R₃ andR₄. Also, if the cut-off frequency ω_(c) for the embodiment of FIG. 4 isto be equal to the cut-off frequency for the embodiment of FIG. 2, thenthe current I in the FIG. 4 embodiment is relatively large as comparedto its signal current i_(s). Hence, the dynamic range of the FIG. 4embodiment is wider than the dynamic range of the FIG. 2 embodiment.

In all of the aforedescribed embodiments, the output voltage V_(out)provided at the emitter electrode of transistor Q₁ is fed backsubstantially in its entirety (that is, approximately 100 percent of theoutput voltage is fed back) to the base electrode of transistor Q₂. Inthe embodiment of FIG. 5, current source A₁ is formed byseries-connected resistors R₅ and R₆ which, as is known, serve as acurrent source if the resistors exhibit a sufficiently high resistancevalue. In addition to functioning as a current source, resistors R₅ andR₆ serve as a voltage divider. Thus, a voltage-divided portion of outputvoltage V_(out) provided at the emitter electrode of transistor Q₁,which transistor is connected in emitter-follower configuration, isapplied to the base electrode of transistor Q₂. The voltage dividingratio K is equal to R₆ /(R₅ +R₆). Accordingly, the voltage which is fedback to the base electrode of transistor Q₂ is equal to KV_(out).

Since only a portion of the output voltage is fed back to the baseelectrode of transistor Q₂, the transfer function H(ω) and the cut-offfrequency ω_(c) of the illustrated filter circuit both are functions ofthis ratio K. The transfer function and the cut-off frequency may,therefore, be expressed as follows: ##EQU6## From equation (11), it isrecognized that, in accordance with the embodiment shown in FIG. 5, thecut-off frequency ω_(c) is reduced by an amount determined by thevoltage-dividing ratio K relative to the embodiment discussed above withrespect to FIG. 2. If, in the FIG. 5 embodiment, the cut-off frequencyis to be equal to that of the FIG. 2 embodiment, then, from equation(11), it is appreciated that current I must be increased. Hence, thedynamic range of the FIG. 5 embodiment will be wider than that of theFIG. 2 embodiment.

FIG. 6 represents a filter circuit which is constructed of two cascadedhigh-pass filter circuits H₁ and H₂ which may be of the constructiondescribed hereinabove with respect to the embodiments of FIGS. 1-5. Inthe filter circuit of FIG. 6, the output terminal of filter H₁ isconnected to the input terminal of filter H₂, and also to an outputterminal T₆. The output terminal of filter H₂ is connected to anamplifier H₃ and also to an output terminal T₇. Amplifier H₃ feeds backa filtered signal to a subtracting circuit H₄, the latter also beingconnected to input terminal T₁ to receive an input signal to befiltered. The output of subtracting circuit H₄, which is equal to thedifference between the input signal and the fed back amplified signal,is supplied to filter H₁ and also to an output terminal T₅.

Let it be assumed that the cut-off frequency of high-pass filter H₁ isless than the cut-off frequency of high-pass filter H₂. If subtractingcircuit H₄ is omitted, for the moment, then the lower frequencies of theinput signal supplied to input terminal T₁ are filtered out by high-passfilter H₁. Thus, a higher frequency signal is passed by filter H₁ tohigh-pass filter H₂. Since the cut-off frequency of filter H₂ is assumedto be higher than that of filter H₁, the lower frequencies of thefiltered signal supplied to filter H₂ are removed. Thus, amplifier H₃ issupplied with a higher frequency signal, the original lower frequenciesof which having been filtered out.

Now, when subtracting circuit H₄ is considered, it is appreciated thatthe higher frequency signal which is provided at the output of amplifierH₃ is subtracted from the original input signal supplied to inputterminal T₁. Thus, subtracting circuit H₄ effectively subtracts, orcancels, the higher frequency signals from the original input signal.Consequently, output terminal T₅ is provided with a low frequency signalwhich, of course, is the difference signal between the input signal andthe higher frequency signal that is subtracted from the input signal.

High-pass filter H₁ removes the lower frequency from the signal providedat the output of subtracting circuit H₄. Thus, output terminal T₆ isprovided with a lower frequency signal whose lower frequencies areremoved, thereby resulting in a band-pass filtered signal.

Thus, the filter circuit illustrated in FIG. 6 functions both as alow-pass filter to supply a low-pass filtered signal at output terminalT₅ ; and also as a band-pass filter to supply a bandpass filtered signalat output terminal T₆.

In another embodiment of the filter circuit in accordance with thepresent invention is illustrated in FIG. 7. This embodiment differs fromthat shown in FIG. 2 primarily in that input terminal T₁ is coupled tothe emitter electrode of transistor Q₁ by a resistor R₇. Also, thecurrent source coupled to the common-connected emitter electrodes oftransistors Q₂ and Q₃, and shown as transistor Q₄ in FIG. 2, is showngenerally as current source A₂ in FIG. 7. Furthermore, current source A₁connected to the emitter electrode of transistor Q₁ in FIG. 2 is shownin FIG. 7 as resistor R₆.

If the output terminal T₂ of the filter circuit shown in FIG. 7 isconnected to the collector electrode of transistor Q₁, it will berecognized that the collector current of this transistor, which is theoutput current I_(out) of the filter circuit, is phase-shifted relativeto the input signal. Assuming that R₆ =R₇, the output current I_(out) isa function of the combination of the signal current i_(s) throughcapacitor C and the current through resistor R₇. The reactance in thecurrent path traversed by signal current i_(s) results in a phase-shiftbetween this current and the current to resistor R₇. Since the signalcurrent i_(s) may be varied by varying the current through currentsource A₂, it is seen that the phase of the output current i_(s) will bevaried as a function of this current source. Thus, the phase of theoutput current may be varied, or controlled, by a suitable controlsignal which determines the current 2I through current source A₂.

More particularly, if R₆ =R₇ =P, and if the emitter voltage oftransistor Q, still is assumed to be V_(out), then the signal (or AC)current flowing through resistor R₆ may be expressed as V_(out) /R.Similarly, the AC current flowing through resistor R₇ may be expressedas ##EQU7## Hence, the total emitter current of transistor Q₁ is##EQU8## This can be rewritten as ##EQU9## Now, ##EQU10## From equations(1) and (2), with the recognition that the signal current i_(s) for theFIG. 7 embodiment is twice the signal current for the FIG. 1 embodimentbecause of the use of the current mirror circuit, the transfer functionH(ω) may be expressed as: ##EQU11## Thus, the total emitter current,which is substantially equal to the collector current I_(out) fortransistor Q₁, is found to be: ##EQU12## The amplitude characteristic ofthe output current I_(out) is, ##EQU13## Thus, the amplitudecharacteristic is constant.

The phase characteristic θ of the output current I_(out) is, ##EQU14##In the foregoing expressions, the term V_(in) /R is equal to the inputcurrent. Since A=1, the output current I_(out) is seen to have the sameamplitude as that of the input current, because ##EQU15## Hence, theoutput current amplitude remains constant even if the current throughcurrent source A₂ varies. But the phase θ of the output current I_(out)is varied relative to the phase of the input current as a function ofthe current I, that is, as a function of the current through currentsource A₂.

Yet another embodiment of the present invention is illustrated in FIG. 8wherein the filter circuit described in FIG. 7 is connected to a crystaloscillator so as to form, in combination therewith, a voltage controlledoscillator (VCO) or, as is sometimes referred to by those of ordinaryskill in the art, a variable crystal oscillator (VXO). Moreparticularly, the output terminal of the phase shift circuit of FIG. 7,that is, the collector electrode of transistor Q₁, is connected to theoscillating circuit formed of transistors Q₇, Q₈ and Q₉, and crystalelement X. Transistors Q₇ and Q₈ are connected in differential amplifierconfiguration. The emitter electrodes of these transistors are connectedin common to a current source A₄. The base electrode of transistor Q₇ isconnected to receive a bias potential equal to the bias potential at thebase electrode of transistor Q₈, thereby maintaining the differentialamplifier in balance. The base electrode of transistor Q₈ is connectedto the collector electrode of transistor Q₁ to receive a phase-shiftedvoltage therefrom corresponding to the aforedescribed output voltageI_(out). The output from this differential amplifier configuration isderived from the collector electrode of transistor Q₈ and is supplied,via emitter-follower transistor Q₉, to crystal element X. The output ofthis crystal element is AC coupled via a capacitor to the input terminalof the phase shift circuit, that is, to capacitor C. The output terminalT₂ of the variable crystal oscillator is connected to the emitterelectrode of transistor Q₉.

It is seen that current source A₂ in FIG. 7 is constituted by transistorQ₄ in FIG. 8, the base electrode of this transistor being coupled to acontrol input T₄ to receive a control signal. The resistor R₈ in thecollector circuit of transistor Q₁ (R₈ =R₆) provides a phase-shiftedvoltage at the collector electrode of transistor Q₁, which phase shiftis a function of the control signal applied to terminal T₄. This hasbeen discussed in detail above.

Crystal element X is of the type which produces an oscillating signal ofa frequency that is determined by the phase of the signal suppliedthereto. That is, if the oscillating signal produced by crystal elementX is fed back thereto through a controllable phase shift circuit, thefrequency of the oscillating signal will vary with the phase shift.Further, if an AC signal whose frequency is equal to the centerfrequency of the crystal element is applied thereto, the oscillatingsignal produced by the crystal element in response thereto will be ofthe center frequency and will be in phase (i.e. zero phase shift) withthe applied AC signal. If the input signal frequency increases above thecenter frequency, the phase of the output oscillating signal isadvanced; and, conversely, if the input signal frequency decreases belowthe center frequency, the phase of the output oscillating signal isretarded.

Now, the capacitor through which the oscillating signal produced bycrystal element X is supplied to the input of the phase shift circuit,imparts a phase advance of +90°. If the control signal applied toterminal T₄ produces a phase shift of -90°, the +90° phase advance iscancelled and the input and output signals of crystal element X are inphase with each other. Hence, the crystal oscillator produces anoscillating signal at the center frequency of crystal element X. If thecontrol signal at terminal T₄ produces a phase shift of -75°, the phaseof the AC signal supplied to the input of crystal element X relative tothe phase of the oscillating signal at its output, is equal to+90°-75°=+15°. This means that the output signal is phase-shifted by-15° relative to the input signal of the crystal element. Consequently,the frequency of the oscillating signal produced by crystal element X isreduced from the center frequency, and this reduced frequency signal atthe output of the crystal element is in phase with the reduced frequencysignal at the input thereof. Conversely, if the control signal atterminal T₄ produces a phase shift of -105°, the phase of the AC signalsupplied to the input of crystal element X relative to the phase of theoscillating signal at its output, is equal to +90°-105°=-15°. Thisresults in an output signal that is phase shifted by +15° relative tothe input signal of the crystal element. Therefore, the frequency of theoscillating signal produced by crystal element X is increased over thecenter frequency.

Control of the phase shift circuit, and thus, the frequency of thecrystal oscillator, is attained by the control signal applied toterminal T₄. The controlled, variable frequency oscillating signal isderived at output terminal T₂ which, it is seen, is connected to theinput of crystal element X. Of course, this output terminal may beconnected to other locations in the illustrated circuit, as desired.

Referring now to FIG. 9, there is illustrated a modification of theembodiment described above with respect to FIG. 1. In this modifiedembodiment, the base electrode of transistor Q₃ is connected to an inputterminal T₈ for receiving an input signal V₂. This differs from thepreviously described embodiment wherein the base electrode of transistorQ₃ is supplied with a bias voltage V_(BB). If input terminal T₁ issupplied with the input signal V₁, then the signal path traversed bysignal current i_(s) is formed of input terminal T₁, capacitor C and theemitter resistances r_(e) of transistors Q₂ and Q₃ to input terminal T₈.The signal current i_(s) through capacitor C is equal to the signalcurrent i_(s) through the emitter resistances of transistors Q₂ and Q₃and may be expressed as: ##EQU16## The terms in equation (12) can berearranged so as to solve for the output voltage V_(out) as follows:##EQU17## wherein R=2kT/q.

The first term in equation (14) represents that the filter circuit shownin FIG. 9 exhibits high-pass characteristics in response to the inputvoltage V₁ supplied to input terminal T₁. The second term in equation(14) represents that the filter circuit exhibits low-passcharacteristics in response to the input voltage V₂ supplied to inputterminal T₈. The cut-off frequency for the high-pass characteristic isequal to the cut-off frequency for the low-pass characteristic. Thesecut-off frequencies are controlled as a function of the current 2Ithrough current source A₂.

If it is assumed that V₁ =-V₂, then the transfer function H(ω) for thefilter circuit shown in FIG. 9 may be expressed as: ##EQU18## Thetransfer function of equation (15) is seen to have a variable phase thatis a function of the current I. Thus, the embodiment of FIG. 9 functionsas a controllable phase shift circuit when V₁ =-V₂ the amount of phaseshift being controlled by the current 2I through current source A₂.

The embodiment of FIG. 9 can be combined with a crystal oscillator ofthe type described previously with respect to FIG. 8 and also can becombined with another, similar phase shift circuit so as to form a huecontrol circuit which finds ready application in a color televisionreceiver. Such a hue control circuit is illustrated in FIG. 10. In thisembodiment, current source A₂ is constituted by current sourcetransistor Q₄ whose base electrode is supplied with an automatic phasecontrol (APC) signal derived from an APC circuit A₁₀. Furthermore,current source A₃, shown in FIG. 9, is constituted by the current mirrorcircuit formed of transistors Q₅ and Q₆, described above with respect tothe embodiment of FIG. 2. The output of the phase shift circuit, thatis, the emitter electrode of transistor Q₁, is connected to the crystaloscillator formed of transistors Q₇ -Q₉ and crystal element X. In thiscrystal oscillator, the base electrode of transistor Q₈ is supplied witha bias voltage and, in this regard, differs from the crystal oscillatorshown in FIG. 8 wherein the base electrode of transistor Q₈ is connectedto the collector electrode of transistor Q₁.

The oscillating signal produced by crystal element X is AC coupled tothe base electrode of a phase-splitting transistor Q₁₀. This transistorhas its collector electrode connected to power supply terminal T₃ by aresistor R₉ and its emitter electrode connected to ground by a resistorR₁₀. The emitter electrode of transistor Q₁₀ is connected to the baseelectrode of transistor Q₃, and the collector electrode of transistorQ₁₀ is connected to capacitor C₁. Hence, it is appreciated that thevoltages which are supplied to capacitor C₁ and to transistor Q₃, thatis, voltages V₁ and V₂ (discussed above with respect to the embodimentof FIG. 9) are equal and opposite to each other. Thus, the circuitformed of transistors Q₁ -Q₆ functions as a controllable phase shiftcircuit similar to that described in FIG. 9, and the amount of phaseshift imparted by this circuit is determined by the APC voltage appliedto current source transistor Q₄. As mentioned with respect to theembodiment of FIG. 8, the frequency of the oscillating signal producedby the crystal oscillator is controlled in accordance with the amount ofphase shift imparted by the phase shift circuit. That is, the frequencyof the oscillating signal provided at the output of crystal element X iscontrolled by the APC voltage applied to transistor Q₄ which, in turn,controls the phase shift imparted by the phase shift circuit.

Transistors Q₁₁ -Q₁₆ are connected in a manner similar to that oftransistors Q₁ -Q₆ and, therefore, form another phase shift circuit. Thecapacitor C₂ in this phase shift circuit is connected to the collectorelectrode of transistor Q₁₀, and the base electrode of transistor Q₁₃ isconnected to the emitter electrode of transistor Q₁₀. Consequently, inthe example assumed hereinabove, the voltages which are supplied tocapacitor C₂ and to transistor Q₁₃ are similar to voltages V₁ and V₂ ofFIG. 9, and are equal but opposite to each other. The output of thissecond phase shift circuit is derived from the emitter electrode oftransistor Q₁₁ and is supplied as a reference signal to a demodulatorA₁₁. The demodulator is included in a typical color television receiver.

A variable resistor R₁₁ is connected to the base electrode of transistorQ₁₄ to supply a base control signal thereto as a function of the settingof the variable resistor. This setting may be adjusted, as desired, bythe viewer of the color television receiver.

In operation, the APC voltage provided by APC circuit A₁₀ isrepresentative of the phase differential between the local oscillatingsignal normally provided in the color television receiver and thereceived burst signal, as is conventional. This APC voltage varies thephase shift imparted by the first phase shift circuit (formed oftransistors Q₁ -Q₆) so as to adjust the frequency of the oscillatingsignal generated by the crystal oscillator. Since the combination ofthis phase shift circuit and the crystal oscillator functions as avariable crystal oscillator (VXO), it is appreciated that the phase ofthe oscillating signal generated by the VXO is locked to the burstsignal by the APC voltage. This phase-locked oscillating signal issupplied, via the second phase shift circuit, to the color demodulatorA₁₁. Since the phase of the oscillating signal which is supplied as areference signal to the color demodulator may be varied as a function ofthe setting of variable resistor R₁₁, it will be recognized that the hueof the reproduced video picture may be varied, as desired, by the viewerbecause it is the phase of the reference signal which determines the hueof the reproduced video picture.

It is appreciated that each of the filter circuits shown in theforegoing embodiments may be constructed as an integrated circuit. As istypical, an IC must be provided with external connecting terminals, orpads, by which it can be electrically connected to other circuitry.These connecting terminals have been shown, in the illustratedembodiments, as terminals T₁, T₂ . . . . The cost of construction of anIC is determined, to a large part, by the number of such externalconnecting terminals that must be provided. In the foregoingembodiments, the filter circuit is shown with a minimum number ofconnecting terminals. Hence, the present invention is readily adaptedfor low cost IC manufacturing.

While the present invention has been particularly shown and describedwith reference to various embodiments, it should be readily appreciatedto those of ordinary skill in the art that various changes andmodifications in form and detail may be made without departing from thespirit and scope of the invention. For example, although the reactancedevice used herein has been shown as a capacitor, it may, alternatively,be replaced by an inductor. If an inductive reactance device is used,the filter circuit exhibits low-pass characteristics.

Also, it is appreciated that, in the various embodiments describedabove, current source A₃ may be replaced by current mirror transistorsQ₅ and Q₆, and vice versa. Furthermore, current source A₁ may be anyconventional current source, such as a current source transistor, arelatively high resistance, or the like. Preferably, current source A₂is formed as a current source transistor whose collector-emitter currentis controlled by a suitable control signal.

It is intended that the foregoing, as well as various other changes andmodifications, be included within the scope of the appended claims.

What is claimed is:
 1. A filter circuit, comprising an input terminal for receiving an input signal; first transistor means having base, emitter and collector electrodes; reactance means connecting the base electrode of said first transistor means to said input terminal; second and third transistor means, each having base, emitter and collector electrodes, said second and third transistor means being connected in differential amplifier configuration and having their emitter electrodes coupled to a common connection; n diodes connected between the emitter electrode of each of said second and third transistor means and said common connection, respectively, current source means connected to said common connection; said first transistor means having its emitter electrode connected to the base electrode of said second transistor means and its base electrode connected to the collector electrode of said second transistor means; an additional current source connected to said emitter electrode of said first transistor means; and an output terminal connected to at least one of the collector and emitter electrodes of said first transistor means to provide an output signal.
 2. The circuit of claim 1 wherein said reactance means comprises a capacitor.
 3. The circuit of claim 1 or 2 wherein said first-mentioned current source means comprises a variable current source, whereby the cut-off frequency of said filter circuit is a function of the current produced by said current source.
 4. The circuit of claim 3 further comprising a second additional current source connected to the collector electrode of said second transistor means.
 5. The circuit of claim 4 further comprising a bias potential supplied to the base electrode of said third transistor means; and a source of operating potential coupled to the collector electrodes of said first and third transistor means.
 6. The circuit of claim 1, wherein said output terminal is connected to the emitter electrode of said first transistor means. 